Transcript
LT1795 Dual 500mA/50MHz Current Feedback Line Driver Amplifier DESCRIPTIO
FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■
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The LT®1795 is a dual current feedback amplifier with high output current and excellent large signal characteristics. The combination of high slew rate, 500mA output drive and up to ±15V operation enables the device to deliver significant power at frequencies in the 1MHz to 2MHz range. Short-circuit protection and thermal shutdown insure the device’s ruggedness. The LT1795 is stable with large capacitive loads and can easily supply the large currents required by the capacitive loading. A shutdown feature switches the device into a high impedance, low current mode, reducing power dissipation when the device is not in use. For lower bandwidth applications, the supply current can be reduced with a single external resistor.
500mA Output Drive Current 50MHz Bandwidth, AV = 2, RL = 25Ω 900V/µs Slew Rate, AV = 2, RL = 25Ω Low Distortion: –75dBc at 1MHz High Input Impedance, 10MΩ Wide Supply Range, ±5V to ±15V Full Rate, Downstream ADSL Supported Low Power Shutdown Mode Power Saving Adjustable Supply Current Stable with CL = 10,000pF Power Enhanced Small Footprint Packages TSSOP-20, S0-20 Wide Available in a 20-Lead TSSOP Package
U APPLICATIO S ■ ■ ■ ■ ■
The LT1795 comes in the very small, thermally enhanced, 20-lead TSSOP package for maximum port density in line driver applications.
ADSL HDSL2, G.lite Drivers Buffers Test Equipment Amplifiers Video Amplifiers Cable Drivers
, LTC and LT are registered trademarks of Linear Technology Corporation.
TYPICAL APPLICATION U
Low Loss, High Power Central Office ADSL Line Driver V+ +IN
+ 1/2 LT1795
12.5Ω
– 1k 1:2* 165Ω
100Ω 1k
– 1/2 LT1795 –IN
12.5Ω
+ V–
* MIDCOM 50215 OR EQUIVALENT
1795 TA01
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LT1795
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ABSOLUTE
AXI U RATI GS
(Note 1)
Supply Voltage ...................................................... ±18V Input Current ...................................................... ±15mA Output Short-Circuit Duration (Note 2) ............ Indefinite Operating Temperature Range ................ – 40°C to 85°C
Specified Temperature Range (Note 3) ... – 40°C to 85°C Junction Temperature ........................................... 150°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C
U W U PACKAGE/ORDER I FOR ATIO TOP VIEW V–
1
20
V–
NC 2
19 NC
–IN 3
18 OUT
+IN 4
17
ORDER PART NUMBER
V+
LT1795CFE LT1795IFE
COMP 1 V+ 2 OUT 3 V–
4
20 COMP 19 V +
LT1795CSW LT1795ISW
18 OUT 17
V–
SHDN 5
16 COMP
V– 5
16 V –
SHDNREF 6
15 COMP
V–
6
15 V –
+IN 7
14 V +
V–
7
14 V –
–IN 8
13 OUT
–IN 8
13 –IN
NC 9
12 NC
+IN 9
12 +IN
V – 10
11 V –
SHDN 10
FE PACKAGE 20-LEAD PLASTIC TSSOP
ORDER PART NUMBER
TOP VIEW
11 SHDNREF
S PACKAGE 20-LEAD PLASTIC SW TJMAX = 150° C, θJA ≈ 40°C/W (Note 4)
TJMAX = 150° C, θJA = 40°C/W (Note 4) UNDERSIDE METAL INTERNALLY CONNECTED TO V– (PCB CONNECTION OPTIONAL)
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full specified temperature range, otherwise specifications are at TA = 25°C. VCM = 0V, ±5V ≤ VS ≤ ±15V, pulse tested, VSHDN = 2.5V, VSHDNREF = 0V unless otherwise noted. (Note 3) SYMBOL
PARAMETER
VOS
Input Offset Voltage
CONDITIONS
MIN
TYP
MAX
UNITS
●
±3 ±4.5
±13 ±17
mV mV
●
±1 ±1.5
±3.5 ±5.0
mV mV
●
10
●
±2 ±8
±5 ±20
µA µA
●
±0.5 ±1.5
±2 ±7
µA µA
●
±10 ±20
±70 ±100
µA µA
●
±10 ±20
±30 ±50
µA µA
Input Offset Voltage Matching Input Offset Voltage Drift IIN
+
Noninverting Input Current Noninverting Input Current Matching
IIN–
Inverting Input Current Inverting Input Current Matching
µV/°C
en
Input Noise Voltage Density
f = 10kHz, RF =1k, RG = 10Ω, RS = 0Ω
3.6
nV/√Hz
+ in
Input Noise Current Density
f = 10kHz, RF =1k, RG = 10Ω, RS = 10kΩ
2
pA/√Hz
– in
Input Noise Current Density
f = 10kHz, RF =1k, RG = 10Ω, RS = 10kΩ
30
pA/√Hz 1795fa
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ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full specified temperature range, otherwise specifications are at TA = 25°C. VCM = 0V, ±5V ≤ VS ≤ ±15V, pulse tested, VSHDN = 2.5V, VSHDNREF = 0V unless otherwise noted. (Note 3) SYMBOL RIN
+
PARAMETER
CONDITIONS
Input Resistance
VIN = ±12V, VS = ±15V V = ±2V, VS = ±5V
MIN
TYP
MAX
UNITS
● ●
1.5 0.5
10 5
MΩ MΩ
Input Capacitance
VIN = ±15V
2
pF
Input Voltage Range (Note 5)
VS = ±15V VS = ±5V
● ●
±12 ±2
±13.5 ±3.5
V V
Common Mode Rejection Ratio
VS = ±15V, VCM = ±12V VS = ±5V, VCM = ±2V
● ●
55 50
62 60
dB dB
Inverting Input Current Common Mode Rejection
VS = ±15V, VCM = ±12V VS = ±5V, VCM = ±2V
● ●
Power Supply Rejection Ratio
VS = ±5V to ±15V
●
Noninverting Input Current Power Supply Rejection
VS = ±5V to ±15V
●
30
500
nA/V
Inverting Input Current Power Supply Rejection
VS = ±5V to ±15V
●
1
5
µA/V
AV
Large-Signal Voltage Gain
VS = ±15V, VOUT = ±10V, RL = 25Ω VS = ±5V, VOUT = ±2V, RL = 12Ω
● ●
55 55
68 68
dB dB
ROL
Transresistance, ∆VOUT/∆IIN –
VS = ±15V, VOUT = ±10V, RL = 25Ω VS = ±5V, VOUT = ±2V, RL = 12Ω
● ●
75 75
200 200
kΩ kΩ
VOUT
Maximum Output Voltage Swing
VS = ±15V, RL = 25Ω ●
±11.5 ±10.0
±12.5 ±11.5
V V
●
±2.5 ±2.0
±3 ±3
V V
●
0.5
CIN+
CMRR
PSRR
VS = ±5V, RL = 12Ω IOUT
Maximum Output Current
VS = ±15V, RL = 1Ω
IS
Supply Current Per Amplifier
VS = ±15V, VSHDN = 2.5V
Supply Current Per Amplifier, RSHDN = 51k, (Note 6)
VS = ±15V
1 1 60
77
VS = ±15V, VSHDN = 0.4V VS = ±15V, VSHDN = 0.4V
●
dB
1
A 34 42
mA mA
15
20 25
mA mA
●
Output Leakage Current, Shutdown
µA/V µA/V
29 ●
Positive Supply Current, Shutdown
10 10
1
200
µA
1
200
µA
Channel Separation
VS = ±15V, VOUT = ±10V, RL = 25Ω
HD2, HD3
2nd and 3rd Harmonic Distortion Differential Mode
f = 1MHz, VO = 20VP-P, RL = 50, AV = 2
SR
Slew Rate (Note 7)
AV = 4, RL = 400Ω
Slew Rate
AV = 4, RL = 25Ω
900
V/µs
Small-Signal BW
AV = 2, VS = ±15V, Peaking ≤ 1.5dB RF = RG = 910Ω, RL = 100Ω
65
MHz
AV = 2, VS = ±15V, Peaking ≤ 1.5dB RF = RG = 820Ω, RL = 25Ω
50
MHz
BW
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Applies to short-circuits to ground only. A short-circuit between the output and either supply may permanently damage the part when operated on supplies greater than ±10V. Note 3: The LT1795C is guaranteed to meet specified performance from 0°C to 70°C and is designed, characterized and expected to meet these extended temperature limits, but is not tested at – 40°C and 85°C. The LT1795I is guaranteed to meet the extended temperature limits.
80
400
110
dB
–75
dBc
900
V/µs
Note 4: Thermal resistance varies depending upon the amount of PC board metal attached to the device. If the maximum dissipation of the package is exceeded, the device will go into thermal shutdown and be protected. Note 5: Guaranteed by the CMRR tests. Note 6: RSHDN is connected between the SHDN pin and V +. Note 7: Slew rate is measured at ±5V on a ±10V output signal while operating on ±15V supplies with RF = 1k, RG = 333Ω (AV = +4) and RL = 400Ω. 1795fa
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RSD = 0Ω, IS = 30mA per Amplifer, VS = ±15V, Peaking ≤ 1dB, RL = 25Ω
RSD = 51kΩ, IS = 15mA per Amplifer, VS = ±15V, Peaking ≤ 1dB, RL = 25Ω
AV
RF
RG
–3dB BW (MHz)
AV
RF
RG
–3dB BW (MHz)
–1
976
976
44
–1
976
976
30
1
1.15k
—
53
1
1.15k
—
32
2
976
976
48
2
976
976
32
10
649
72
46
10
649
72
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TYPICAL PERFOR A CE CHARACTERISTICS
V+
VS = ±15V AV = 1 RL = ∞ RSD = 0Ω
30 25 20 15
RSD = 51kΩ
10 5 0 –50 –25
50 25 75 0 TEMPERATURE (°C)
100
VS = ±15V
–1
2.0 RL = 2k
–2
RL = 25Ω
–3 –4
4 3
RL = 25Ω
2 1
RL = 2k
V– –50 –25
125
25 50 0 75 TEMPERATURE (°C)
LT1795 G01
DISTORTION (dBc)
CURRENT INTO SHDN PIN (mA)
–50
0.3 0.2
1 2 3 4 VOLTAGE APPLIED AT SHDN PIN (V)
SOURCING
1.2 SINKING
1.0 0.8
50 25 75 0 TEMPERATURE (°C)
100
–60
5
1795 G04
125
LT1795 G03
Third Harmonic Distortion vs Frequency –40
AV = 2 DIFFERENTIAL VOUT = 20VP-P VS = ±15V RLOAD = 50Ω IQ PER AMPLIFIER
–50
IQ = 5mA
–70 IQ = 10mA –80 –90
0.1
0
1.4
0.6 –50 –25
125
–40
0.4
0
1.6
Second Harmonic Distortion vs Frequency
VS = ±15V VSHDNREF = 0V
0.5
100
VS = ±15V
1.8
LT1795 G02
SHDN Pin Current vs Voltage 0.6
Output Short-Circuit Current vs Junction Temperature
DISTORTION (dBc)
35
OUTPUT SATURATION VOLTAGE (V)
SUPPLY CURRENT PER AMPLIFIER (mA)
40
Output Saturation Voltage vs Junction Temperature OUTPUT SHORT-CIRCUIT CURRENT (A)
Supply Current vs Ambient Temperature
–100 10k
IQ = 15mA
–70 –80
IQ = 5mA
IQ = 10mA IQ = 20mA
–90
IQ = 20mA 100k FREQUENCY (Hz)
–60
AV = 2 DIFFERENTIAL VOUT = 20VP-P VS = ±15V RLOAD = 50Ω IQ PER AMPLIFIER
1M LT1795 G05
IQ = 15mA –100 10k
100k FREQUENCY (Hz)
1M LT1795 G06
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TYPICAL PERFOR A CE CHARACTERISTICS Second Harmonic Distortion vs Frequency
Third Harmonic Distortion vs Frequency
–40
–60
AV = 10 DIFFERENTIAL VOUT = 20VP-P VS = ±15V RLOAD = 50Ω IQ PER AMPLIFIER
–70
IQ = 5mA
–50
IQ = 20mA
–70 –80
IQ = 10mA
–50
DISTORTION (dBc)
–60
–40
–40 AV = 10 DIFFERENTIAL VOUT = 20VP-P VS = ±15V RLOAD = 50Ω IQ PER AMPLIFIER
DISTORTION (dBc)
DISTORTION (dBc)
–50
Second Harmonic Distortion vs Frequency
IQ = 10mA
–80
IQ = 15mA –90
IQ = 5mA
IQ = 20mA
–90
–60
AV = 2 DIFFERENTIAL VOUT = 20VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER IQ = 5mA
–70 IQ = 10mA –80
IQ = 15mA
–90
IQ = 15mA –100 10k
100k FREQUENCY (Hz)
–100 10k
1M
100k FREQUENCY (Hz)
1M
LT1795 G07
Third Harmonic Distortion vs Frequency –40
–40
IQ = 5mA –70 IQ = 20mA –80
–60
AV = 10 DIFFERENTIAL VOUT = 20VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
–70
AV = 10 DIFFERENTIAL VOUT = 20VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
–50
IQ = 20mA
–80
IQ = 10mA
DISTORTION (dBc)
–50
DISTORTION (dBc)
DISTORTION (dBc)
–60
–60
IQ = 5mA
–70 –80
IQ = 15mA –90
–90
IQ = 15mA
IQ = 10mA –100 10k
100k FREQUENCY (Hz)
–100 10k
1M
100k FREQUENCY (Hz)
–80 –90
–40
IQ = 10mA IQ = 20mA
IQ = 15mA
–60
AV = 2 DIFFERENTIAL VOUT = 4VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
–50
IQ = 5mA
DISTORTION (dBc)
DISTORTION (dBc)
DISTORTION (dBc)
–50
IQ = 5mA
1M
Second Harmonic Distortion vs Frequency
–40
–70
100k FREQUENCY (Hz)
LT1795 G12
Third Harmonic Distortion vs Frequency
–40
–60
IQ = 15mA
–100 10k
1M
IQ = 20mA
LT1795 G11
Second Harmonic Distortion vs Frequency
–50
IQ = 10mA
–90
IQ = 5mA
LT1795 G10
AV = 2 DIFFERENTIAL VOUT = 4VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
1M LT1795 G09
Second Harmonic Distortion vs Frequency
–40 –50
100k FREQUENCY (Hz)
LT1795 G08
Third Harmonic Distortion vs Frequency AV = 2 DIFFERENTIAL VOUT = 20VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
IQ = 20mA
–100 10k
IQ = 10mA –70 –80
IQ = 15mA
–90
–60
AV = 10 DIFFERENTIAL VOUT = 4VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER IQ = 10mA
–70 –80 –90
IQ = 5mA
IQ = 15mA
IQ = 20mA
IQ = 20mA
–100 10k
100k FREQUENCY (Hz)
1M LT1795 G13
–100 10k
100k FREQUENCY (Hz)
1M LT1795 G14
–100 10k
100k FREQUENCY (Hz)
1M LT1795 G15
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TYPICAL PERFOR A CE CHARACTERISTICS Third Harmonic Distortion vs Frequency –40
IQ = 5mA
IQ = 10mA
–70
IQ = 15mA
–80 –90
–60
IQ = 5mA IQ = 10mA
–80
IQ = 20mA
IQ = 20mA
–90
100k FREQUENCY (Hz)
1M
–50
–70
–100 –110 10k
–40 AV = 2 DIFFERENTIAL VOUT = 4VP-P VS = ±5V RLOAD = 50Ω IQ PER AMPLIFIER
–50 DISTORTION (dBc)
DISTORTION (dBc)
–60
AV = 10 DIFFERENTIAL VOUT = 4VP-P VS = ±12V RLOAD = 50Ω IQ PER AMPLIFIER
–100 10k
100k FREQUENCY (Hz)
1M
–40
DISTORTION (dBc)
DISTORTION (dBc)
–50
IQ = 20mA IQ = 15mA IQ = 10mA
–60
–100 10k
–100 10k
AV = 10 DIFFERENTIAL VOUT = 4VP-P VS = ±5V RLOAD = 50Ω IQ PER AMPLIFIER
1M
100k FREQUENCY (Hz)
1M LT1795 G20
–3dB Bandwidth vs Supply Current 50
1000 800 FALLING 600
VS = ±15V TA =25°C AV = 4 RLOAD = 25Ω RF = 1k 25 15 20 30 7.5 10 SUPPLY CURRENT PER AMPLIFIER (mA) 1795 • G21
–3dB BANDWIDTH (MHz)
45
RISING SLEW RATE (V/µs)
IQ = 10mA
IQ = 20mA
–100 10k
1200
0
IQ = 5mA
–90
100k FREQUENCY (Hz)
1M
IQ = 15mA
–80
Slew Rate vs Supply Current
200
100k FREQUENCY (Hz)
LT1795 G18
LT1795 G19
400
IQ = 20mA
–70
IQ = 5mA
–90
IQ = 15mA
–80
Third Harmonic Distortion vs Frequency
AV = 10 DIFFERENTIAL VOUT = 4VP-P VS = ±5V RLOAD = 50Ω IQ PER AMPLIFIER
–80
IQ = 10mA
–70
LT1795 G17
–40
–70
IQ = 5mA
–90
Second Harmonic Distortion vs Frequency
–60
–60
AV = 2 DIFFERENTIAL VOUT = 4VP-P VS = ±5V RLOAD = 50Ω IQ PER AMPLIFIER
IQ = 15mA
LT1795 G16
–50
DISTORTION (dBc)
–40 –50
Third Harmonic Distortion vs Frequency
Second Harmonic Distortion vs Frequency
40
35
30
25
VS = ±15V TA =25°C AV = 4 RLOAD = 25Ω RF = 1k
25 15 20 30 7.5 10 SUPPLY CURRENT PER AMPLIFIER (mA) 1795 • G22
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The LT1795 is a dual current feedback amplifier with high output current drive capability. The amplifier is designed to drive low impedance loads such as twisted-pair transmission lines with excellent linearity.
When VSHDN = VSHDNREF, the device is shut down. The device will interface directly with 3V or 5V CMOS logic when SHDNREF is grounded and the control signal is applied to the SHDN pin. Switching time between the active and shutdown states is about 1.5µs.
SHUTDOWN/CURRENT SET
Figures 1 to 4 illustrate how the SHDN and SHDNREF pins can be used to reduce the amplifier quiescent current. In both cases, an external resistor is used to set the current. The two approaches are equivalent, however the required resistor values are different. The quiescent current will be approximately 115 times the current in the SHDN pin and 230 times the current in the SHDNREF pin. The voltage across the resistor in either condition is V + – 1.5V. For example, a 50k resistor between V + and SHDN will set the
If the shutdown/current set feature is not used, connect SHDN to V + and SHDNREF to ground. The SHDN and SHDNREF pins control the biasing of the two amplifiers. The pins can be used to either turn off the amplifiers completely, reducing the quiescent current to less then 200µA, or to control the quiescent current in normal operation.
V+ V+
10 SHDN
RSHDN
11 SHDNREF 10 SHDN
RSHDNREF
11 SHDNREF
1795 F03
1795 F01
Figure 1. RSHDN Connected Between V + and SHDN (Pin 10); SHDNREF (Pin 11) = GND. See Figure 2
Figure 3. RSHDNREF Connected Between SHDNREF (Pin 11) and GND; SHDN (Pin 10) = V +. See Figure 4
80
80 TA = 25°C VS = ±15V
70 AMPLIFIER SUPPLY CURRENT, ISY – mA (BOTH AMPLIFIERS)
AMPLIFIER SUPPLY CURRENT, ISY – mA (BOTH AMPLIFIERS)
70 60 50 40 30 20 10
TA = 25°C VS = ±15V
60 50 40 30 20 10
0
0 0
25
50
75 100 125 150 175 200 225 RSHDN (kΩ) 1795 F02
Figure 2. LT1795 Amplifier Supply Current vs RSHDN. RSHDN Connected Between V+ and SHDN, SHDNREF = GND (See Figure 1)
50 100 150 200 250 300 350 400 450 500 RSHDNREF (kΩ) 1795 F04
Figure 4. LT1795 Amplifier Supply Current vs RSHDNREF. RSHDNREF Connected Between SHDNREF and GND, SHDN = V+ (See Figure 3)
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quiescent current to 33mA with VS = ±15V. If ON/OFF control is desired in addition to reduced quiescent current, then the circuits in Figures 5 to 7 can be employed.
Figure 8 illustrates a partial shutdown with direct logic control. By keeping the output stage slightly biased on, the output impedance remains low, preserving the line termination. The design equations are:
V+
R1 =
RSHDN RB 10k
OFF ON
10 SHDN Q1
INTERNAL LOGIC THRESHOLD ~1.4V
11 SHDNREF
115 • VH
(IS )ON – (IS )OFF
(0V)
(
(3.3V/5V) Q1: 2N3904 OR EQUIVALENT
R2 =
1795 F05
Figure 5. Setting Amplifier Supply Current Level with ON/OFF Control, Version 1
RPULLUP >500k
OFF (0V) (3.3V/5V)
RSHDN2 RB2 10k
ON Q1A
OFF (0V) (3.3V/5V)
Q1B
VCC
Q1A, Q1B: ROHM IMX1 or FMG4A (W/INTERNAL RB)
OFF (0V) IPROG (3.3V/5V) IPROG ≅ 0.5mA FOR REXT = 0Ω (SEE SHDN PIN CURRENT vs VOLTAGE CHARACTERISTIC)
ON OFF (0V) (3.3V/5V)
Figure 6. Setting Multiple Amplifier Supply Current Levels with ON/OFF Control, Version 2
REXT
VSHDN = Shutdown Pin Voltage ≈1.4V VCC = Positive Supply Voltage
1795 F06
ON
(IS)ON = Supply Current Fully On (IS)OFF = Supply Current Partially On
11 SHDNREF
RB1 10k
(VSHDN / VH ) • (IS )ON – (IS )OFF + (IS )OFF
VH = Logic High Level 10 SHDN
ON
)
where
V+
RSHDN1
115 • VCC – VSHDN
R2 R1 10 SHDN
INTERNAL LOGIC THRESHOLD ISY ~ 1.4V CONTROL
11 SHDNREF
1795 F08
Figure 8. Partial Shutdown
SHDN 10 ISY CONTROL
INTERNAL LOGIC THRESHOLD ~ 1.4V
SHDNREF 11 1795 F07
Figure 7. Setting Amplifier Supply Current Level with ON/OFF Control, Version 3
THERMAL CONSIDERATIONS The LT1795 contains a thermal shutdown feature that protects against excessive internal (junction) temperature. If the junction temperature of the device exceeds the protection threshold, the device will begin cycling between normal operation and an off state. The cycling is not harmful to the part. The thermal cycling occurs at a slow rate, typically 10ms to several seconds, which depends on the power dissipation and the thermal time constants of the package and heat sinking. Raising the ambient tempera1795fa
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ture until the device begins thermal shutdown gives a good indication of how much margin there is in the thermal design. For surface mount devices, heat sinking is accomplished by using the heat spreading capabilities of the PC board and its copper traces. For the TSSOP package, power is dissipated through the exposed heatsink. For the SO package, power is dissipated from the package primarily through the V – pins (4 to 7 and 14 to 17). These pins should have a good thermal connection to a copper plane, either by direct contact or by plated through holes. The copper plane may be an internal or external layer. The thermal resistance, junction-to-ambient will depend on the total copper area connected to the device. For example, the thermal resistance of the LT1795 connected to a 2 × 2 inch, double sided 2 oz copper plane is 40°C/W. CALCULATING JUNCTION TEMPERATURE The junction temperature can be calculated from the equation: TJ = (PD)(θJA) + TA where TJ = Junction Temperature TA = Ambient Temperature PD = Device Dissipation θJA = Thermal Resistance (Junction-to-Ambient) Differential Input Signal Swing The differential input swing is limited to about ±5V by an ESD protection device connected between the inputs. In normal operation, the differential voltage between the input pins is small, so this clamp has no effect. However, in the shutdown mode, the differential swing can be the same as the input swing. The clamp voltage will then set the maximum allowable input voltage.
ing 0.5A current peaks into the load, a 1Ω power supply impedance will cause a droop of 0.5V, reducing the available output swing by that amount. Surface mount tantalum and ceramic capacitors make excellent low ESR bypass elements when placed close to the chip. For frequencies above 100kHz, use 1µF and 100nF ceramic capacitors. If significant power must be delivered below 100kHz, capacitive reactance becomes the limiting factor. Larger ceramic or tantalum capacitors, such as 4.7µF, are recommended in place of the 1µF unit mentioned above. Inadequate bypassing is evidenced by reduced output swing and “distorted” clipping effects when the output is driven to the rails. If this is observed, check the supply pins of the device for ripple directly related to the output waveform. Significant supply modulation indicates poor bypassing. Capacitance on the Inverting Input Current feedback amplifiers require resistive feedback from the output to the inverting input for stable operation. Take care to minimize the stray capacitance between the output and the inverting input. Capacitance on the inverting input to ground will cause peaking in the frequency response (and overshoot in the transient response), but it does not degrade the stability of the amplifier. Feedback Resistor Selection The optimum value for the feedback resistors is a function of the operating conditions of the device, the load impedance and the desired flatness of response. The Typical AC Performance tables give the values which result in less than 1dB of peaking for various resistive loads and operating conditions. If this level of flatness is not required, a higher bandwidth can be obtained by use of a lower feedback resistor. For resistive loads, the COMP pin should be left open (see Capacitive Loads section). Capacitive Loads
POWER SUPPLY BYPASSING To obtain the maximum output and the minimum distortion from the LT1795, the power supply rails should be well bypassed. For example, with the output stage supply-
The LT1795 includes an optional compensation network for driving capacitive loads. This network eliminates most of the output stage peaking associated with capacitive loads, allowing the frequency response to be flattened. 1795fa
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W
U
APPLICATIONS INFORMATION Figure 9 shows the effect of the network on a 200pF load. Without the optional compensation, there is a 6dB peak at 85MHz caused by the effect of the capacitance on the output stage. Adding a 0.01µF bypass capacitor between the output and the COMP pins connects the compensation 14
VS = ±15V CL = 200pF
12
RF = 3.4k NO COMPENSATION
VOLTAGE GAIN (dB)
10 RF = 1k COMPENSATION
8 6 4 2
and greatly reduces the peaking. A lower value feedback resistor can now be used, resulting in a response which is flat to ±1dB to 45MHz. The network has the greatest effect for CL in the range of 0pF to 1000pF. Although the optional compensation works well with capacitive loads, it simply reduces the bandwidth when it is connected with resistive loads. For instance, with a 25Ω load, the bandwidth drops from 48MHz to 32MHz when the compensation is connected. Hence, the compensation was made optional. To disconnect the optional compensation, leave the COMP pin open.
0 –2
–6
DEMO BOARD
RF = 3.4k COMPENSATION
–4 1
10 FREQUENCY (MHz)
A demo board (DC261A) is available for evaluating the performence of the LT1795. The board is configured as a differential line driver/receiver suitable for xDSL applications. For details, consult your local sales representative.
100 1795 F09
Figure 9
U
PACKAGE DESCRIPTIO
SW Package 20-Lead Plastic Small Outline (Wide .300 Inch) (Reference LTC DWG # 05-08-1620) .050 BSC .045 ±.005
.030 ±.005 TYP
.496 – .512 (12.598 – 13.005) NOTE 4
N 20
18
17
16
15
14
13
12
11
N
.325 ±.005
.420 MIN
19
.394 – .419 (10.007 – 10.643)
NOTE 3
1
2
3
N/2 N/2
RECOMMENDED SOLDER PAD LAYOUT
.005 (0.127) RAD MIN
.009 – .013 (0.229 – 0.330) NOTE: 1. DIMENSIONS IN
.291 – .299 (7.391 – 7.595) NOTE 4 .010 – .029 × 45° (0.254 – 0.737)
1
2
3
4
5
6
7
8
.093 – .104 (2.362 – 2.642)
9
10 .037 – .045 (0.940 – 1.143)
0° – 8° TYP
NOTE 3 .016 – .050 (0.406 – 1.270)
.050 (1.270) BSC .014 – .019 (0.356 – 0.482) TYP
.004 – .012 (0.102 – 0.305) S20 (WIDE) 0502
INCHES (MILLIMETERS) 2. DRAWING NOT TO SCALE 3. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS. THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS 4. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
1795fa
10
LT1795 U
PACKAGE DESCRIPTIO
FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663)
Exposed Pad Variation CA
6.40 – 6.60* (.252 – .260) 4.95 (.195)
4.95 (.195)
20 1918 17 16 15 14 13 12 11
6.60 ±0.10
2.74 (.108)
4.50 ±0.10
2.74 6.40 (.108) BSC
SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 9 10
RECOMMENDED SOLDER PAD LAYOUT
1.20 (.047) MAX
4.30 – 4.50* (.169 – .177) 0° – 8°
0.09 – 0.20 (.0036 – .0079)
0.45 – 0.75 (.018 – .030)
NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE
0.65 (.0256) BSC
0.195 – 0.30 (.0077 – .0118)
0.05 – 0.15 (.002 – .006) FE20 (CA) TSSOP 0203
4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE
1795fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LT1795 W W SI PLIFIED SCHEMATIC SHDN
TO ALL CURRENT SOURCES
SHDNREF
V+
Q5
Q10
Q2
D1 Q6
Q1
Q11 Q15
Q9
V– +IN
CC
–IN
V–
50Ω COMP
RC
OUTPUT V+ V+
Q12
Q3
Q8
Q16
Q14
D2
Q4 Q7
Q13
V– 1795 SS
RELATED PARTS PART NUMBER
DESCRIPTION
COMMENTS
LT1497
Dual 125mA, 50MHz Current Feedback Amplifier
900V/µs Slew Rate
LT1207
Dual 250mA, 60MHz Current Feedback Amplifier
Shutdown/Current Set Function
LT1886
Dual 200mA, 700MHz Voltage Feedback Amplifier
Low Distortion: –72dBc at 200kHz
1795fa
12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT/TP 0603 1K REVA • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1999
