Transcript
VMMK-3213 Using the VMMK-3213 Wafer Scale Packaged Detector in 6 to 18 GHz Applications
Application Note 5526
Introduction The VMMK-3213 is a broadband directional coupler with integrated temperature compensated detector designed for 6 to 18 GHz applications. The detector provides a DC output proportional to RF power input, providing a means of measuring amplifier power output. The VMMK-3213 is a three-terminal device with the “through” 50 Ω transmission line connecting directly between the RF input and RF output ports. A DC bias is fed to the RF input port and the rectified DC is available at the RF output port.
Using the VMMK-3213 With only three terminals available, the DC bias and detected voltage are internally DC coupled to the input and output terminals respectively. The key to successful operation of the VMMK-3213 is the use of low loss bias decoupling networks connected to both the RF input port and the RF output port. A simple circuit is shown in Figure 1. The bias decoupling networks are very similar to those used to bias up a discrete transistor. Both networks provide a low loss AC coupled RF path to the device and a means of DC biasing the device on the input and a
means of extracting the detected voltage on the output of the device. Bias decoupling networks in the 6 to 18 GHz frequency range usually consist of quarter wave high impedance lines followed by low impedance quarter wave stubs. These are inherently narrow band compared to the operating frequency range of the VMMK-3213. Adding some resistance in series, in the form of R1 and R2, may enhance bandwidth. The internal load resistor for the detector is approximately 20 kΩ. If desired, resistor R3 can be used as an external load resistor for the detector. Although C4 provides additional decoupling, any shunt capacitance at C4 will reduce the video bandwidth of the detector and therefore may not be desirable. The -3 dB video bandwidth of the detector itself is approximately 30 MHz. Any additional bypassing external to the device at the output terminal will decrease the bandwidth. More detailed information will follow in a later section of this application note. The suggested bias voltage at the RF input port is 1.5 V. At this nominal bias, the bias current is typically 0.16 mA. With no RF input power a nominal 60 mV Voffset voltage appears at the detected output port. A plot of DC output versus RF input for the VMMK-3213 is shown in Figure 2. 3.5
C2
RF Input
RF Output
R2
R1 R3
C3
C4
Output DC Voltage (V)
C1
6 GHz 12 GHz 18 GHz
3
VMMK-3213
2.5 2 1.5 1 0.5 0 -5
Vb
Vdet
Figure 1. Biasing the VMMK-3213 detector module
0
5
10 15 Input Power (dBm)
20
25
Figure 2. Output DC voltage vs. RF input power for the VMMK-3213
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PCB Pattern
Demonstrating Performance
Implementing the bias networks is usually done in microstrip. The recommended printed circuit board via pattern is shown in Figure 3. This is a non-solder mask defined footprint (NSMD). The outline of the solder mask that borders the device is shown by the area indicated in green. The recommended footprint does not require any plated through holes under the device. Modeling and tests indicate that placing vias adjacent to (within 0.003”) and on either side of the device, as shown in Figure 3, provides good grounding for the VMMK-3XXX series devices when mounted on 0.010” thick RO4350 printed circuit board material.
Demonstrating performance with a demonstration board requires mounting the VMMK-3213 on a 50 Ω microstrip line with connectors. Rogers 4350 printed circuit board material with a 10 mil thickness is used as a low loss substrate for launching in and out of the VMMK-3213. The 50 Ω line width is 0.020”. The printed circuit board stack is a multi-layer stack which provides rigidity during testing. The total thickness is 0.060”. A Johnson SMA connector, part number 142-0761-861, is used to provide a smooth transition to the microstrip line. Bias decoupling networks have been included on the demonstration board to inject a voltage at the input port and as a means to measure the detected voltage at the output port.
Additional information covering the assembly, cleaning and handling of VMMK products is covered in Avago Application Note AN-5378.
1.2 (0.048) 0.400 (0.016)
0.100 (0.004)
To demonstrate the loss of the VMMK-3213 by itself requires de-embedding all of the printed circuit board losses, including the 50 Ω microstrip line, the bias decoupling lines and the connectors. A completed demonstration board for the VMMK-3213 is shown in Figure 4.
0.100 (0.004)
0.500 (0.020)
0.500 (0.020) 0.200 (0.008)
Part of Input Circuit
Part of Output Circuit
0.200 (0.008)
0.076 max (0.003) 2 pl see discussion 0.381 (0.015) 2 pl
0.254 dia PTH (0.010) 4 pl Solder Mask 0.400 dia (0.016) 4 pl
Figure 3. Recommended PCB layout for VMMK devices
Figure 4. VMMK-3213 demonstration board
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Measuring the loss of any low loss component over a wide range of frequencies when it is embedded within a circuit board is difficult. The quarter wave bias decoupling lines and open circuit trapezoidal stubs provide a low loss means of providing bias decoupling; however, they are inherently narrow band.
0.7 (0.028)
A reference demonstration board including all microstrip lines and bias networks but without the VMMK-3213 was built. The difference in loss between the reference board and the demonstration board shown in Figure 4 is the loss of the VMMK-3213 by itself. A parts list is shown in Figure 5. The demonstration board design center frequency is 10 GHz. The main frequency limiting structures are the quarter wave bias decoupling lines and trapezoidal capacitors. This design can be optimized for other frequencies in the 6 to 18 GHz frequency range. The nominal value for C1 and C2 is 1 pF.
Component
Description
C1, C2
1 pF see text (ATC 600 series)
C3
1000 pF
C4
Not used
U1
VMMK-3213
R1, R2
100 Ω (not critical, see text)
R3
Not used (optional, see text)
Figure 5. VMMK-3213 demonstration board parts list
The response of the demonstration board and reference boards is shown in Figure 6. The red curve represents the loss of a straight through piece of 50 Ω microstrip line the same length as the demonstration board. This response also includes the effect of the SMA connectors. The blue curve represents the loss of the bias decoupling lines and all circuitry except the VMMK-3213. The green curve represents the entire demonstration board including the VMMK-3213. Subtracting the green curve from the blue curve will suggest the loss of just the VMMK-3213. Note that the lowest loss occurs at 10 GHz, which is the center frequency of the bias decoupling networks. At 10 GHz the additional loss due to the VMMK-3213 is less than 0.3 dB. Although Figure 6 shows increased loss at 8 and 12 GHz compared to 10 GHz, the increase in loss is due primarily to increased mismatch due to inadequate bandwidth of the bias decoupling networks used on the demonstration board. More elaborate bias decoupling networks could be used if needed, but for typical narrow band applications requiring less than 10% bandwidth, the quarter-wave bias decoupling networks should be adequate. Another factor that can influence the performance of the VMMK-3213 is the inductance provided by the printed circuit board vias. In general, the lowest inductance will provide the lowest insertion loss and best return loss performance of the VMMK-3213. The effect of the printed circuit board vias is best evaluated using a linear simulator such as Agilent’s Advanced Design System and the published data sheet S-parameters for the VMMK-3213. The recommended PCB layout, as shown in Figure 3, should minimize the inductance associated with the vias.
S-parameter Measurements The VMMK-3213 device S-parameters are measured on a 0.016 inch thick RO4003 printed circuit test board, using 300 μm pitch G-S-G (ground signal ground) probes. A coplanar waveguide is used to provide a smooth transition from the probes to the device under test. The presence of the ground plane on top of the test board results in excellent grounding at the device under test. A combination of SOLT (Short – Open – Load – Thru) and TRL (Thru - Reflect - Line) calibration techniques are used to correct for the effects of the test board, resulting in accurate device S-parameters.
Other circuit approaches The VMMK-3213 data sheet discusses a coplanar waveguide (CPW) approach (Figure 7) that may integrate into a customer’s application better than microstrip line. Although a CPW approach can provide a lower inductance path for grounding the common lead of the VMMK-3213, producing other circuit elements such as couplers and bias networks may be more difficult to model. The CPW approach is built on 0.016” thick RO4003. The 50 Ω transmission line is 0.020” in width which matches up well with the input pad of the VMMK-3213. A 0.005” gap between the transmission line and the top side CPW ground plane provides optimum dimensions for a 50 Ω transmission line.
0
dB (s(16,15)) dB (s(14,13)) dB (s(12,11))
-1
-2
-3
-4 4
6
8
10 freq. GHz
12
14
15
Figure 6. Insertion loss vs frequency response of VMMK-3213 demonstration board (green) vs, reference demonstration board (blue). The difference in the curves is the loss of the VMMK-3213. A through (red) with 50 Ω microstrip line is also shown for reference.
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Figure 7. CPW approach for the demonstration board
Biasing the device on the CPW board is accomplished by using a small 0402 resistor in place of the quarter wave bias decoupling lines used in the microstrip line approach. A 22 kΩ resistor can be used to bias the VMMK-3213 from a 5 V power supply. With the nominal 0.16 mA of device current required by the VMMK-3213, the bias voltage at the device will be a nominal 1.5 V. Finding the lowest parasitic and lowest loss resistor may require some empirical bench testing where various component losses are measured by shunting each component across the 50 Ω CPW line and measuring its effect on the loss of the CPW transmission line. The resistor can be bypassed with a small 1 pF capacitor in parallel with a 1000 pF capacitor.
pulse handling capability of the VMMK-3213 similar to its end use, a suggestion would be to use two coaxial to waveguide transitions back to back in series with the RF output 50 Ω termination. The back-to-back transitions will provide a low loss high pass filter which will minimize the effect of the 50 Ω termination on pulse handling capability. The length of cable between the RF output of the demonstration board and the load should be minimized because a 50 Ω coaxial line will add approximately 30 pF/ft.
In a similar fashion, a single resistor can be used on the output to extract the detected voltage, Vdet from the output port. A smaller 1 kΩ value resistor may be more appropriate as any additional capacitive bypassing after the 1 kΩ resistor will further decrease the video bandwidth of the detector.
In systems where the VMMK-3213 is followed by something other than an antenna or waveguide probe which is normally a DC open, it may be beneficial to use quarter wave length parallel coupled microstrip lines to AC couple to the following stage. The quarter wave parallel coupled lines will provide a low loss DC block at the design center frequency with less series capacitance than a typical capacitor. Depending on what follows the DC block, its effect of loading the Vdet port may be less than if a capacitor was used.
Video Bandwidth
Coupler Directivity
The -3 dB video bandwidth of the VMMK-3213 by itself is approximately 30 MHz and is based on the expression:
The directivity of a coupler describes its ability to differentiate between the power being sent in one direction down a transmission line versus power being sent in the opposite direction. The couplers used in laboratory test equipment are usually in the vicinity of 40 dB which are considered very good. The requirements on couplers used to measure forward and reverse power as a check on system operability are normally not as stringent. The VMMK-3213 has a typical directivity of 15 dB at 6 and 12 GHz increasing to 17 dB at 18 GHz.
fu (3 dB) = 0.25 / tw to 0.35 / tw tw = the pulse width The VMMK-3213 by itself will pass pulses as narrow as 10 ns. Since the detected voltage at the output port is resident at the same terminal as the RF output, anything attached to either the Vdet port or the RF output port of the demonstration board will restrict video bandwidth. The easiest solution on the Vdet port is to not use any bypass capacitance at C4 and keep R2 as low a value as possible. Using a scope probe with any appreciable capacitance will also adversely effect video bandwidth and pulse rise time. If possible use a high impedance low capacitance operational amplifier as a load for the Vdet port. In a normal application of the VMMK-3213, the RF output is an antenna which presents a DC open to the device. When testing the VMMK-3213 in a demonstration board the normal load that a piece of test equipment would present to the device would be a nominal resistive 50 Ω load. In the case of the VMMK-3213 demonstration board, the external 50 Ω RF port termination is AC coupled with a 1 pF blocking capacitor shown as C2. The series load of 1 pF and 50 Ω is in shunt with the Vdet voltage and will severely limit the ability of the VMMK-3213 to demodulate narrow pulses. The demonstration board was designed to evaluate the loss performance of the VMMK-3213 and to provide a DC voltage proportional to a continuous wave signal present at the RF input. In order to evaluate the
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Directivity is measured by inserting the VMMK-3213 between a well matched 50 Ω signal source and a well match 50 Ω termination. Incident power is set for a convenient Vdet level. As an example, it could be set for 200 mV. The VMMK-3213 is then reversed and the incident power increased until the same 200 mV is achieved. The difference in the incident power levels is the directivity of the VMMK-3213 at that particular frequency. Coupler directivity can be correlated to an error coefficient that can affect the magnitude of the power detected by the coupler. Since we are only making scalar measurements without any phase information, we can only calculate a plus/minus window that our measured coupled power could be found. This can happen with two loads with the same return loss but with different phase angles which can cause two different levels of power being coupled to the detector. The greater the reflection from the load, the greater the error could be. The potential error Δρ relates to coupler directivity by the following equation: Δρ = A + A ρL2
The reflection coefficient of the output load is ρL and A = Log10-1 (directivity (dB)/-20).
Table 1. Uncertainty vs. directivity of the VMMK-3213 Directivity (dB)
RL (dB)
ρL
Δρ
ML ± (dB)
The following equation relates ρ to mismatch loss:
15
-18
0.126
0.181
± 0.14
Mismatch loss (dB) = -10 log10 (1-ρ2)
-12
0.251
0.189
± 0.16
Mismatch in dB suggests a plus and minus window over which the detected power could vary based on the reflection coefficient of the load attached to the coupler. Table 1 shows a few scenarios describing the mismatch loss versus several different return loss loads and 15 dB and 17 dB coupler directivity.
-6
0.501
0.223
± 0.22
-18
0.126
0.143
± 0.09
-12
0.251
0.150
± 0.10
-6
0.501
0.176
± 0.14
The data shows that for greater than a 12 dB return loss load, the maximum uncertainty in measured power for a 17 dB directivity coupler is plus or minus 0.1 dB. The uncertainty increases to plus or minus 0.16 dB for a 15 dB directivity coupler.
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Summary The VMMK-3213 provides an integrated directional coupler and temperature compensated detector that provides a compact approach for monitoring amplifier power output. Its useful frequency range is from 6 to 18 GHz.
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Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies in the United States and other countries. Data subject to change. Copyright © 2005-2011 Avago Technologies. All rights reserved. AV02-3048EN - July 13, 2011
